ASmartSolid-StateFuseCuresBoost-ConverterIlls

The majority of asynchronous, inductor-based boost converters (the step-up switching types) exhibit a DC-current path between power source and load (Figure 1). This path can have two undesirable consequences: First, if a grounded output or other overload draws heavy output current for more than a few hundred milliseconds, the catch diode (usually a Schottky type) can exude that blended aroma of moen silon and potting compound familiar to all true hackers. Second, if switching action is disabled for any reason, such as intentional shutdown, the load voltage remains just src="/data/attachment/portal/201007/ET12535201007191111331.gif">
  Figure 1. The problem of a direct path for current flow from source to load is intrinsic to the boost-converter topology.
  
  Both problems are neatly solved for the relatively low output-current applications (<5A) that employ monolithic current-mode controllers and high-side current sensing. These circuits replace the catch diode with a synchronous switching transistor that can be disabled by shutdown or removal of input power. Disabling this internal transistor or turning it off during shutdown removes the path for DC current flow. The load then sees a requisite high-impedance disconnect. When not in shutdown, the circuit's cycle-by-cycle current-sensing mechanism (using an internal, high-side, current-sense resistor) protects against catastrophic meltdown from internal current overloads. Finally, thermal-overload protection provides a safe area of operation (SAO).
  
  For applications with higher output current, in which pricing makes synchronous switching impractical for monolithic devices, the load-disconnect function requires a high-side switch external to the controller die. A discrete current-mode topology using a high-side current-sense resistor and a synchronous switching transistor is possible, but that approach suffers from PC-board parasitics and layout dependencies, especially at high switching frequencies. The result is a relatively complex design, particularly when system constraints mandate a low input voltage (<3.6V).
  
  A synchronous, high-side external switch becomes feasible at higher levels of peak inductor current (>5A), but cost and complexity override heat and efficiency considerations at the more moderate levels of inductor current (~1.5A to 5A) discussed in this article. A simple catch diode is again the most desirable solution. The challenge is to achieve the desired load disconnect while retaining use of the humble catch diode and the unadorned boost topology.
  
  A simple and smart solution is presented in Figure 2, where a MAX668 controller illustrates the demanding task of boosting from low input voltages. This current-mode boost controller drives a logic-level n-channel enhancement-mode MOSFET (configured src="/data/attachment/portal/201007/ET12535201007191111332.gif">
  Figure 2. This boost converter with load disconnect illustrates the minimum-cost configuration.
  
  The key element in implementing a smart load disconnect is the p-channel enhancement-mode MOSFET Q1. As shown, the system can enable this boost circuit (active-low src="/data/attachment/portal/201007/ET12535201007191111333.gif">
  Figure 3. Adding three components in the MOSFET-gate circuit of Figure 2 provides the slow-connect/fast-disconnect necessary to accommodate heavy startup loads.
  
  These circuits require a logic-level p-channel MOSFET such as Q1 to fully enhance the channel and obtain a low Rds-on. If Q1's src="/data/attachment/portal/201007/ET12535201007191111334.gif">
  Figure 4. Further embellishments (to Figure 3) add remote-sense regulation and low-voltage detection to the boost converter with load disconnect.
  
  The MAX4544 operates within data-sheet limits for supply voltages as low as 2.7V. With the input supply at 3.3V and about 0.3V across the Schottky diode, the MAX4544 (and MAX810) remains operational even when the boost converter is shut down. The MAX810 output is high during shutdown, connecting the MAX4544 COM node to NO (the Q1 source). When the boost converter turns src="/data/attachment/portal/201007/ET12535201007191111335.gif">
  Figure 5. To achieve the load disconnect for low output voltages, this circuit ensures adequate drive to the MOSFET by generating a negative rail for the gate-drive IC (MAX810L).
  
  Although Q1 turns src="/data/attachment/portal/201007/ET12535201007191111336.gif">
  Figure 6. Extending the load-disconnect idea to non-boost-converter circuits forms a solid-state fuse that is applicable to any DC-power bus.
  
  Suppose, for example, that -48 volts are to be protected against overcurrent. We interrupt the rail side instead of the ground side, because the voltage source is negative and we use an n-channel FET plus a MAX809T reset circuit, whose reset-output polarity is opposite to that of the MAX810. The supply voltage can range down to -36V under normal operation (Figure 7).
  
  
  Figure 7. This solid-state fuse protects a negative DC-power bus.
  
  Design equations are as follows:
  
  The MAX809 quiescent current is about 100&micro;A maximum over temperature, and the current through Rh and RL should be about 100 times' higher to minimize the effect of quiescent current on trip voltage: 36/(Rh+RL) = 10 mA, therefore
  
  
  
  (Rh+RL) = 3600 Ohms.
  
  The MAX809 threshold is much lower than the supply-trip voltage, so RL is smaller than Rh, approximately by the ratio Vthreshold/(Vthreshold + Vsupply-trip) = 3/(36+3) = 0.077. Thus, MAX809 Iq flows through ~93.3% of (Rh+RL), causing a voltage-trip contribution of ~0.336V. Taking this fact into account, set the initial trip voltage for calculating Rh and RL at 36V - 0.336V = 35.664V. Using 1% resistors for Rh and RL, Vsupply-trip = 35.664V. This threshold occurs when the MAX809T threshold is at its minimum (3.15V, over the temperature range -°C to + 85°C):
  
  
  
  35.664V[RL(0.99)/(RL(0.99)+Rh(1.01))] = 3.15V.
  
  Calculated values for RL and Rh are 323.81Ω and 3276.19Ω, respectively. The closest 1% values are 320 and 3280. Considering these resistor values and the 100&micro;A Iq, the maximum supply-trip voltage becomes 36.09V, which surpasses 36V slightly. This result also occurs only for simultaneous worst-case values for all errors, which are rare scenarios in practice. For most applications, the foregoing design would be quite acceptable. The MAX809's nominal threshold voltage gives a nominal trip voltage of -34.65V.
  
  Rh should have a power rating of 0.5W. Because the voltage across RL exceeds the MAX809's maximum input-voltage rating when Vsupply goes above its minimum limit, place a 5V ±5% zener diode across RL as shown.